While it is difficult for base stations to estimate the millimeter wave (mmWave) channels and find the optimal mmWave beam for user equipments (UEs) quickly, the sub-6 GHz channels which are usually easier to obtain and more robust to blockages could be used to reduce the time before initial access and enhance the reliability of mmWave communication. Considering that the channel information is collected by a massive number of radio base stations and would be sensitive to privacy and security, Federated Learning (FL) is a match for this use case. In practice, the channel vectors are usually subject to Non-Independently Distributed (non-IID) distributions due to the greatly varying wireless communication environments between different radio base stations and their UEs. To achieve satisfying performance for all radio base stations instead of only the majority of them, a useful solution is designing personalized methods for each radio base station. In this thesis, we implement two personalized FL methods including 1) Finetuning FL Model on Private Dataset of Each Client and 2) Adaptive Expert Models for FL to predict the optimal mmWave beamforming vector directly from the non-IID sub-6 GHz channel vectors generated from DeepMIMO. According to our experimental results, Finetuning FL Model on Private Dataset of Each Client achieves higher average mmWave downlink spectral efficiency than the global FL. Besides, in terms of the average Top-1 and Top-3 classification accuracies, its performance improvement over the global FL model even exceeds the improvement of the global FL over the pure local models. / Även om det är svårt för en basstation att uppskatta en kanal för millimetervåg (mmWave) och snabbt hitta den bästa mmWave-strålen för en användarutrustning (UE), kan den dra fördel av kanaler under 6 GHz, som i allmänhet är mer lättillgängliga och mer motståndskraftig mot blockering, för att minska tid för första besök och förbättra tillförlitligheten hos mmWave-kommunikation. Med tanke på att kanalinformation samlas in av ett stort antal radiobasstationer och är känslig för integritet och säkerhet är federated learning (FL) väl lämpat för detta användningsfall. I praktiken, eftersom den trådlösa kommunikationsmiljön varierar mycket mellan olika radiobasstationer och deras UE, följer kanalvektorer vanligtvis en icke-oberoende distribution (icke-IID). För att uppnå tillfredsställande prestanda för alla radiobasstationer, inte bara de flesta radiobasstationer, är en användbar lösning att utforma ett individuellt tillvägagångssätt för varje radiobasstation. I detta dokument implementerar vi två personliga FL-metoder, inklusive 1) finjustering av FL-modellen på varje klients privata datauppsättning och 2) en adaptiv expertmodell av FL för att direkt generera icke-IID sub-6 GHz kanalvektorer förutsäga optimal mmWave beamforming vektorer. Enligt våra experimentella resultat uppnår finjustering av FL-modellen på varje klients privata datauppsättning högre genomsnittlig mmWave-nedlänksspektral effektivitet än global FL. Dessutom överträffar dess prestandaförbättring jämfört med den globala FL-modellen till och med den för den globala FL jämfört med den rent lokala modellen vad gäller genomsnittlig klassificeringsnoggrannhet i topp-1 och topp-3.

No description available.

Interest in millimeter (mm-) wave frequencies covering the spectrum of 30-300 GHz has been steadily increasing. Advantages such as larger absolute bandwidth and smaller form-factor have made this frequency region attractive for numerous applications, including high-speed wireless communication, sensing, material science, health, automotive radar, and space exploration. Continuous development of silicon-germanium heterojunction bipolar transistor (SiGe HBT) and associated BiCMOS technology has achieved transistors with fT/fmax of 505/720 GHz and integration with 55 nm CMOS. Such accomplishment and predictions of beyond THz performance have made SiGe BiCMOS technology the most competitive candidate for addressing the aforementioned applications. Especially for mobile applications, a critical demand for future mm-wave applications will be low DC power consumption (Pdc), which requires a substantial reduction of supply voltage and current. Conventionally, reducing the supply voltage will lead to HBTs operating close to or in the saturation region, which is typically avoided in mm-wave circuits due to expectated performance degradation and often inaccurate models. However, due to only moderate speed reduction at the forward-biased base-collector voltage (VBC) up to 0.5 V and the accuracy of the compact model HICUM/L2 also in saturation, low-power mm-wave circuits with SiGe HBTs operating in saturation offer intriguing benefits, which have been explored in this thesis based on 130 nm SiGe BiCMOS technologies: • Different low-power mm-wave circuit blocks are discussed in detail, including low-noise amplifiers (LNAs), down-conversion mixers, and various frequency multipliers covering a wide frequency range from V-band (50-75 GHz) to G-band (140-220 GHz). • Aiming at realizing a better trade-off between Pdc and RF performance, a drastic decrease in supply voltage is realized with forward-biased VBC, forcing transistors of the circuits to operate in saturation. • Discussions contain the theoretical analysis of the key figure of merits (FoMs), topology and bias selection, device sizing, and performance enhancement techniques. • A 173-207 GHz low-power amplifier with 23 dB gain and 3.2 mW Pdc, and a 72-108 GHz low-power tunable amplifier with 10-23 dB gain and 4-21 mW Pdc were designed. • A 97 GHz low-power down-conversion mixer was presented with 9.6 dB conversion gain (CG) and 12 mW Pdc. • For multipliers, a 56-66 GHz low-power frequency quadrupler with -3.6 dB peak CG and 12 mW Pdc, and a 172-201 GHz low-power frequency tripler with -4 dB peak CG and 10.5 mW Pdc were realized. By cascading these two circuits, also a 176-193 GHz low-power ×12 multiplier was designed, achieving -11 dBm output power with only 26 mW Pdc. • An integrated 190 GHz low-power receiver was designed as one receiving channel of a G-band frequency extender specifically for a VNA-based measurement system. Another goal of this receiver is to explore the lowest possible Pdc while keeping its highly competitive RF performance for general applications requiring a wide LO tuning range. Apart from the low-power design method of circuit blocks, the careful analysis and distribution of the receiver FoMs are also applied for further reduction of the overall Pdc. Along this line, this receiver achieved a peak CG of 49 dB with a 14 dB tunning range, consuming only 29 mW static Pdc for the core part and 171 mW overall Pdc, including the LO chain. • All designs presented in this thesis were fabricated and characterized on-wafer. Thanks to the accurate compact model HICUM/L2, first-pass access was achieved for all circuits, and simulation results show excellent agreement with measurements. • Compared with recently published work, most of the designs in this thesis show extremely low Pdc with highly competitive key FoMs regarding gain, bandwidth, and noise figure. • The observed excellent measurement-simulation agreement enables the sensitivity analysis of each design for obtaining a deeper insight into the impact of transistor-related physical effects on critical circuit performance parameters. Such studies provide meaningful feedback for process improvement and modeling development.:Table of Contents Kurzfassung . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ii Abstract . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . iv Table of Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . vii 1 Introduction 1 1.1 Motivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 1.2 Objectives . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 List of symbols and acronyms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 2 Technology 7 2.1 Fabrication Technologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 2.1.1 SiGe HBT performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 2.1.2 B11HFC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 2.1.3 SG13G2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 2.1.4 SG13D7 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 2.2 Commonly Used Components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 2.2.1 Grounded-sidewall-shielded microstrip line . . . . . . . . . . . . . . . . . . 12 2.2.2 Zero-impedance Transmission Line . . . . . . . . . . . . . . . . . . . . . . 15 2.2.3 Balun . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 2.2.3.1 Active Balun . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 2.2.3.2 Passive Balun . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 2.3 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 3 Low-power Low-noise Amplifiers 25 3.1 173-207 GHz Ultra-low-power Amplifier . . . . . . . . . . . . . . . . . . . . . . . 25 3.1.1 Topology Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 3.1.2 Bias Dependency of the Small-signal Performance . . . . . . . . . . . . . 27 3.1.2.1 Bias . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 3.1.2.2 Bias vs Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 3.1.2.3 Bias vs Noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 3.1.2.4 Bias vs Stability . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 3.1.3 Bias selection and Device sizing . . . . . . . . . . . . . . . . . . . . . . . . 36 3.1.3.1 Bias Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 3.1.3.2 Device Sizing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37 3.1.4 Performance Enhancement Technologies . . . . . . . . . . . . . . . . . . . 41 3.1.4.1 Gm-boosting Inductors . . . . . . . . . . . . . . . . . . . . . . . 41 3.1.4.2 Stability Enhancement . . . . . . . . . . . . . . . . . . . . . . . 43 3.1.4.3 Noise Improvement . . . . . . . . . . . . . . . . . . . . . . . . . 45 3.1.5 Circuit Realization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 3.1.5.1 Layout Scheme . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 3.1.5.2 Inductors Design . . . . . . . . . . . . . . . . . . . . . . . . . . . 47 3.1.5.3 Dual-band Matching Network . . . . . . . . . . . . . . . . . . . 48 3.1.5.4 Circuit Implementation . . . . . . . . . . . . . . . . . . . . . . . 50 3.1.6 Results and Discussions . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51 3.1.6.1 Measurement Setup . . . . . . . . . . . . . . . . . . . . . . . . . 51 3.1.6.2 Measurement Results . . . . . . . . . . . . . . . . . . . . . . . . 51 3.1.6.3 Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54 3.2 72-108 GHz Low-Power Tunable Amplifier . . . . . . . . . . . . . . . . . . . . . . 55 3.2.1 Configuration, Sizing, and Bias Tuning Range . . . . . . . . . . . . . . . . 55 3.2.2 Regional Matching Network . . . . . . . . . . . . . . . . . . . . . . . . . . 57 3.2.2.1 Impedance Variation . . . . . . . . . . . . . . . . . . . . . . . . . 57 3.2.2.2 Regional Matching Network Design . . . . . . . . . . . . . . . . 60 3.2.3 Circuit Implementation . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65 3.2.4 Results and Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65 3.2.4.1 Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65 3.2.4.2 Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68 3.3 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71 4 Low-power Down-conversion Mixers 73 4.1 97 GHz Low-power Down-conversion Mixer . . . . . . . . . . . . . . . . . . . . . 74 4.1.1 Mixer Design and Implementation . . . . . . . . . . . . . . . . . . . . . . 74 4.1.1.1 Mixer Topology . . . . . . . . . . . . . . . . . . . . . . . . . . . 74 4.1.1.2 Bias Selection and Device Sizing . . . . . . . . . . . . . . . . . . 77 4.1.1.3 Mixer Implementation . . . . . . . . . . . . . . . . . . . . . . . . 79 4.1.2 Results and Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80 4.1.2.1 Measurement Results . . . . . . . . . . . . . . . . . . . . . . . . 80 4.1.2.2 Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82 4.2 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83 5 Low-power Multipliers 87 5.1 General Design Flow . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88 5.2 56-66 GHz Low-power Frequency Quadrupler . . . . . . . . . . . . . . . . . . . . 89 5.3 172-201 GHz Low-power Frequency Tripler . . . . . . . . . . . . . . . . . . . . . 93 5.4 176-193 GHz Low-power ×12 Frequency Multiplier . . . . . . . . . . . . . . . . . 96 5.5 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100 6 Low-power Receivers 101 6.1 Receiver Performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101 6.2 LO Chain (×12) Integrated 190 GHz Low-Power Receiver . . . . . . . . . . . . . 104 6.2.1 Receiver Architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 105 6.2.2 Low-power Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . 107 6.2.3 Building Blocks . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 108 6.2.3.1 LNA and LO DA . . . . . . . . . . . . . . . . . . . . . . . . . . 108 6.2.3.2 Tunable Mixer and IF BA . . . . . . . . . . . . . . . . . . . . . 111 6.2.3.3 65 GHz (V-band) Quadrupler . . . . . . . . . . . . . . . . . . . 116 6.2.3.4 G-band Tripler . . . . . . . . . . . . . . . . . . . . . . . . . . . . 120 6.2.4 Receiver Results and Discussion . . . . . . . . . . . . . . . . . . . . . . . 123 6.2.5 Measurement Setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124 6.2.6 Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124 6.3 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 131 7 Conclusions 133 7.1 Summaries . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 133 7.2 Outlook . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 134 Bibliography 135 List of Figures 149 List of Tables 157 A Derivation of the Gm 159 A.1 Gm of standard cascode stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . 159 A.2 Gm of cascode stage with Lcas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 160 A.3 Gm of cascode stage with Lb . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 161 B Derivation of Yin in the stability analysis 163 C Derivation of Zin and Zout 165 C.1 Zin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 165 C.2 Zout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 167 D Derivation of the cascaded oP1dB 169 E Table of element values for the designed circuits 171

The infrared and millimeter-wave portions of the spectrum both have their advantages for development of imaging systems. Because of the difference in wavelengths, infrared imagers offer inherently high resolution, while millimeter-wave systems have better penetration through atmospheric aerosols such as fog and smoke. Shared-aperture imaging systems employing a common focal-plane array that responds to both wavebands are desirable from the viewpoint of overall size and weight. We have developed antenna-coupled sensors that respond simultaneously at 30 THz and at 94 GHz, utilizing electron-beam lithography. Slot-antenna designs were found to be particularly suitable for coupling radiation into metal-oxide-metal (MOM) tunnel diodes at both frequencies. The MOM diodes are fabricated in a layered structure of Ni-NiO-Ni, and act as rectifying contacts. With contact areas as low as 120 nm × 120 nm, these diodes have time constants commensurate with rectification at frequencies across the desired millimeter-wave and infrared bands. One challenge in the development of true focal-plane array imagers across this factor-of-300 bandwidth is that the optimum spatial sampling interval on the focal plane is different in both bands. We have demonstrated a focal plane with interleaved infrared and millimeter-wave sensors by fabricating infrared antennas in the ground plane of the millimeter-wave antenna. Measured performance data in both bands are presented for individual antenna-coupled sensors as well as for devices in the dual-band focal-plane-array format.

Today’s world is becoming increasingly automated and interconnected with billions of smart devices coming online, leading to a steep rise in energy consumption from small microelectronics. This coincides with an urgent push to transform global energy production to green energies, causing disruptions and energy shortages, and making the case for efficient energy use ever more pressing. Two major areas where high growth is expected are the fields of wireless communication and radar sensors. Millimeter-wave frequency bands are planned for fifth-generation (5G) and sixth-generation (6G) cellular communication standards, as well as automotive frequency-modulated continuous wave (FMCW) radar systems for driving assistance and automation. Fast silicon-based technologies enable these advances by operating at high maximum frequencies, such as the silicon-germanium (SiGe) heterojunction bipolar transistor (HBT) technologies. However, even the fastest transistors suffer from low and energy expensive gains at millimeter-wave frequencies. Rather than incremental improvements in circuit efficiency using conventional approaches, a disruptive revolution for green microelectronics could be enabled by exploring the low-power benefits of the super-regenerative receiver for some applications. The super-regenerative receiver uses a regenerative oscillator circuit to increase the gain by positive feedback, through coupling energy from the output back into the input. Careful bias and control of the circuit enables a very large gain from a small number of transistors and a very low energy dissipation. Thus, the super-regenerative oscillator could be used to replace amplifier circuits in high data rate wireless communication systems, or as active reflectors to increase the range of FMCW radar systems, greatly reducing the power consumption. The work in this thesis presents fundamental scientific research into the topic of energy-efficient millimeter-wave super-regenerative receivers for use in civilian wireless communication and radar applications. This research work covers the theory, analysis, and simulations, all the way up to the proof of concept, hardware realization, and experimental characterization. Analysis and modeling of regenerative oscillator circuits is presented and used to improve the understanding of the circuit operation, as well as design goals according to the specific application needs. Integrated circuits are investigated and characterized as a proof of concept for a high data rate wireless communication system operating between 140–220 GHz, and an automotive radar system operating at 60 GHz. Amplitude and phase regeneration capabilities for complex modulation are investigated, and principles for spectrum characterization are derived. The circuits are designed and fabricated in a 130 nm SiGe HBT technology, combining bipolar and complementary metal-oxide semiconductor (BiCMOS) transistors. To prove the feasibility of the research concepts, the work achieves a wireless communication link at 16 Gbit/s over 20 cm distance with quadrature amplitude modulation (QAM), which is a world record for the highest data rate ever reported in super-regenerative circuits. This was powered by a super-regenerative oscillator circuit operating at 180 GHz and providing 58 dB of gain. Energy efficiency is also considerably high, drawing 8.8 mW of dc power consumption, which corresponds to a highly efficient 0.6 pJ/bit. Packaging and module integration innovations were implemented for the system experiments, and additional broadband circuits were investigated to generate custom quench waveforms to further enhance the data rate. For radar active reflectors, a regenerative gain of 80 dB is achieved at 60 GHz from a single circuit, which is the best in its frequency range, despite a low dc power consumption of 25 mW.

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Modern communication systems face new technological challenges, such as the narrowness and overload of the conventional frequency bands employed for these applications. Nowadays, communication systems are expected to operate at higher frequencies, such as the mm-wave band. In particular, for space applications, specific environmental conditions make it necessary to design low-profile, lightweight and high gain systems with wide-angle scanning capabilities. Traditional solutions are reflectors antennas or planar arrays. Reflectors often end up being bulky, whereas array antennas are lossy and costly. Lens antennas, unpopular at low frequencies due to their large size, offer a better solution in this context, due to their focusing properties, wide-scanning capability, and broadband behaviour. Among lens antennas, geodesic lens antennas have recently increased interest since they are fullymetallic and easy to manufacture. Previous research aiming at reducing the profile of geodesic lens antennas, while preserving high performances, allowed a total height reduction by a factor 4. In this work, I investigate the possibility of reducing the profile even further by following a different approach. Instead of folding by mirroring the curved profile, the lens antenna is built with circular ridge structures, in an attempt to discretize the original profile. Different approaches have been proposed. First, designs with different numbers of squared ridges were proposed. The reflections are reduced by chamfering the corners of the ridges. Moreover, triangle ridges and alternating the ridges orientation have also been investigated. The final design has four squared ridges with the same orientation. This design was chosen due to its radiation performance. This approach reduces the profile by a factor 18. A prototype has been manufactured working on the frequency band [24,34] GHz. The scanning range is ±62◦ , reflections levels are below -15 dB and at 29 GHz the maximum realized gain is equal to 15.75 dBi. This solution offers attractive properties, mainly due to its compactness. The height of the lens antenna is restricted by the flare, which was set at λ/2. This means that this lens antenna can be stacked in a linear array with grating-lobe-free performance in the elevation plane. / Moderna kommunikationssystem står inför nya tekniska utmaningar, såsom smalheten samt överbelastning inom de konventionella frekvensband som avsatts för tillhörande applikationer. Nutida kommunikationssystem förväntas operera på högre frekvenser, vilket implicerar våglängder på millimeternivå. Särskilt inom rymdapplikationer så finns förutbestämda miljömässiga förhållanden som nödvändiggör användning av lågprofilerade och lättviktiga system med hög antennförstärkning samt möjlighet för vidvinkelskanning. Traditionella lösningar omfattar både reflektorantenner och plana gruppantenner, vilket antingen är otympligt respektive kostsamt. Linsantenner, otympliga och därav opopulära val inom lägre frekvenser, visar sig vara bra lösningar i given kontext. Detta följer av linsernas fokuseringsegenskaper, breda skanningsförmåga samt naturligt stora frekvensband. Inom guppen av linsantenner så har geodetiska linsantenner fått ökat intresse till följd av dess simpla tillverkningsprocess samt fullt metalliska struktur. Tidigare forskning som syftat åt att minska profilen tillsammande med bibehållen prestanda, har lyckats minska höjden men en faktor av fyra. I detta arbete så undersöks möjligheten att krympa profilen ytterligare via användning av ett nytt angreppssätt. I stället för att vika linsen överstämmande med en kurvig profil, så formas linsantennerna med cirkulära ås-strukturer (små böjningar) i strävan efter att diskretisera den ursprunglig profilen. Olika tillvägagångssätt visas i detta arbete. Först visas profiler med ett varierande antal kvadratiska åsar. Reflektioner längs profilen reduceras vid introduktion av fasningar av kvadratens tillhörande hörn. Ytterligare så har triangulära åsar samt riktningen (ås med riktning upp eller riktning ned längs den horisontella profilen) av samtliga typer utvärderats. Den slutliga designen har fyra kvadratiska åsar i samma riktning, ett designval baserat på strålningsprestanda. Arbetet visar att det sistnämnde tillvägagångssättet minskar profilen med en faktor av 18. En fungerande prototyp inom frekvensbandet [24,34] GHz har tillverkats baserat på sistnämnd design, som uppnår ett skanningsområde upp till 62◦ , en reflektionsnivå under -15 dB samt en maximal antennförstärkning på 15.75 dBi vid 29 GHz. Den föreslagna lösningen erhåller attraktiva egenskaper, främst med avseende på dess kompakthet. Höjden på linsantennen begränsas av en matchande flank med en halv våglängd stor öppning, så att flertalet linsantenner kan staplas och forma en linjär gruppantenn vars prestanda utesluter större sidolober längs höjdplanet.

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