Die vorliegende Arbeit ist der analytischen Untersuchung und Weiterentwicklung von Methoden und Schaltungen zur Einstellung der Signalphase und -amplitude gewidmet. Hierbei wird zum Ziel gesetzt, die Leistungsfähigkeit dieser Schaltungen als analoge Hochfrequenz-Baugruppen in Empfangs- und Sendeschaltkreisen mit einem vergleichbaren oder geringerem schaltungstechnischen Aufwand und Strombedarf zu verbessern und dies anhand von Implementierungsbeispielen zu bestätigen. Die Dämpfungsglied-Topologien , T, überbrücktes T und X werden modelliert und hinsichtlich der Phasenbeeinflussung analysiert, sodass eine Bewertung ihrer Eignung durchgeführt werden kann. Weiterhin wird ein innovativer Ansatz zur Linearisierung der Steuerkennlinie vorgestellt und mit Hilfe einer Beispielschaltung mit einem Phasenfehler von 3 ° und einem Steuerlinearitätsfehler von 0,35 dB innerhalb der 1 dB Grenzfrequenz und einem Steuerbereich von 20 dB nachgewiesen. Die Arbeit bietet darüber hinaus eine analytische Betrachtung zu aktiven steuerbaren Verstärkern, welche die besondere Eignung der Gilbert-Zelle aufzeigt und eine geeignete Ansteuerschaltung ableitet. Am Beispiel nach diesem Prinzip entworfener Schaltkreise werden Phasenfehler von nur 0,4 ° innerhalb eines besonders hohen Stellbereichs von 36 dB demonstriert, wodurch eine Vergrößerung des Stellbereichs um den Faktor 4 und eine Verbesserung des Phasenfehlers um den Faktor 2 im Vergleich zum Stand der Technik erreicht wurde. Es wird der Zirkulator-Phasenschieber maßgeblich durch eine neuartige geeignete Ansteuerung verbessert. Damit werden die sonst für die Amplitudenbeeinflussung im Wesentlichen verantwortlichen Varaktoren überflüssig, ohne dabei den schaltungstechnischen Aufwand zu erhöhen. Eine Messung der entsprechenden Schaltung bestätigt dies mit einem Amplitudenfehler von nur 0,9 dB für einen Phasenstellbereich von 360 °, was einer Verringerung des Fehlers um den Faktor 3 im Vergleich zu herkömmlichen Zirkulator-Phasenschiebern entspricht. Abschließend wird der Funktionsnachweis mehrerer entworfener Vektor-Modulatoren mit einer effektiven Genauigkeit von bis zu 6 bit in Einzelschaltungen, Hybridaufbauten und schließlich im Rahmen eines vollständig integrierten Empfängerschaltkreises erbracht. Dieser erzielt eine Verdopplung der Reichweite bei einer um nur 35% höheren Leistungsaufnahme gegenüber einem herkömmlichen Kommunikationsverfahren (SISO). / The present work is dedicated to the investigation and enhancement of amplitude and phase control methods and circuits. The aim is to enhance the performance of these circuits in modern radio frequency transceivers with a comparable or even lower effort and power consumption. A prove of concept will be delivered with implementation examples. By means of models of the passive attenuator topologies , T, bridged-T and X, a thorough analysis is performed in order to compare them regarding their impact on the signal phase. Additionally, a novel approach to increase the control linearity of the attenuators is proposed and verified by measurements, showing a phase error of 3 ° and a control linearity error of 0,35 dB at the 1 dB corner frequency, successfully. The work also presents an investigation on variable gain amplifiers and reveals the superior performance of the Gilbert cell with respect to low phase variations. A cascode biasing circuit that supports these properties is proposed. Measurements prove this concept with relative phase errors of 0,4 ° over a wide attenuation control range of 36 dB thus cutting the error by half in a four times wider control range. The circulator based phase shifting approach is chosen and improved significantly by means of tuning the transconductor instead of the varactors thus removing their impact on signal amplitude. The approach is supported by measurements yielding an amplitude error of only 0,9 dB within a phase control range of 360 ° which corresponds to an improvement by a factor of three compared to recent circulator phase shifters. Finally, the design of several vector modulator topologies is shown with hardware examples of single chips, hybrid printed circuit boards and highly integrated system level ICs demonstrating a full receiver. By using improved variable gain amplifiers, an effective vector modulator resolution of 6 bit without calibration is achieved. Furthermore, a multiple-input multiple-output system is demonstrated that doubles the coverage range of common SISO systems with only 35% of additional power consumption.

Interest in millimeter (mm-) wave frequencies covering the spectrum of 30-300 GHz has been steadily increasing. Advantages such as larger absolute bandwidth and smaller form-factor have made this frequency region attractive for numerous applications, including high-speed wireless communication, sensing, material science, health, automotive radar, and space exploration. Continuous development of silicon-germanium heterojunction bipolar transistor (SiGe HBT) and associated BiCMOS technology has achieved transistors with fT/fmax of 505/720 GHz and integration with 55 nm CMOS. Such accomplishment and predictions of beyond THz performance have made SiGe BiCMOS technology the most competitive candidate for addressing the aforementioned applications. Especially for mobile applications, a critical demand for future mm-wave applications will be low DC power consumption (Pdc), which requires a substantial reduction of supply voltage and current. Conventionally, reducing the supply voltage will lead to HBTs operating close to or in the saturation region, which is typically avoided in mm-wave circuits due to expectated performance degradation and often inaccurate models. However, due to only moderate speed reduction at the forward-biased base-collector voltage (VBC) up to 0.5 V and the accuracy of the compact model HICUM/L2 also in saturation, low-power mm-wave circuits with SiGe HBTs operating in saturation offer intriguing benefits, which have been explored in this thesis based on 130 nm SiGe BiCMOS technologies: • Different low-power mm-wave circuit blocks are discussed in detail, including low-noise amplifiers (LNAs), down-conversion mixers, and various frequency multipliers covering a wide frequency range from V-band (50-75 GHz) to G-band (140-220 GHz). • Aiming at realizing a better trade-off between Pdc and RF performance, a drastic decrease in supply voltage is realized with forward-biased VBC, forcing transistors of the circuits to operate in saturation. • Discussions contain the theoretical analysis of the key figure of merits (FoMs), topology and bias selection, device sizing, and performance enhancement techniques. • A 173-207 GHz low-power amplifier with 23 dB gain and 3.2 mW Pdc, and a 72-108 GHz low-power tunable amplifier with 10-23 dB gain and 4-21 mW Pdc were designed. • A 97 GHz low-power down-conversion mixer was presented with 9.6 dB conversion gain (CG) and 12 mW Pdc. • For multipliers, a 56-66 GHz low-power frequency quadrupler with -3.6 dB peak CG and 12 mW Pdc, and a 172-201 GHz low-power frequency tripler with -4 dB peak CG and 10.5 mW Pdc were realized. By cascading these two circuits, also a 176-193 GHz low-power ×12 multiplier was designed, achieving -11 dBm output power with only 26 mW Pdc. • An integrated 190 GHz low-power receiver was designed as one receiving channel of a G-band frequency extender specifically for a VNA-based measurement system. Another goal of this receiver is to explore the lowest possible Pdc while keeping its highly competitive RF performance for general applications requiring a wide LO tuning range. Apart from the low-power design method of circuit blocks, the careful analysis and distribution of the receiver FoMs are also applied for further reduction of the overall Pdc. Along this line, this receiver achieved a peak CG of 49 dB with a 14 dB tunning range, consuming only 29 mW static Pdc for the core part and 171 mW overall Pdc, including the LO chain. • All designs presented in this thesis were fabricated and characterized on-wafer. Thanks to the accurate compact model HICUM/L2, first-pass access was achieved for all circuits, and simulation results show excellent agreement with measurements. • Compared with recently published work, most of the designs in this thesis show extremely low Pdc with highly competitive key FoMs regarding gain, bandwidth, and noise figure. • The observed excellent measurement-simulation agreement enables the sensitivity analysis of each design for obtaining a deeper insight into the impact of transistor-related physical effects on critical circuit performance parameters. Such studies provide meaningful feedback for process improvement and modeling development.:Table of Contents Kurzfassung . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ii Abstract . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . iv Table of Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . vii 1 Introduction 1 1.1 Motivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 1.2 Objectives . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 List of symbols and acronyms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 2 Technology 7 2.1 Fabrication Technologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 2.1.1 SiGe HBT performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 2.1.2 B11HFC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 2.1.3 SG13G2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 2.1.4 SG13D7 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 2.2 Commonly Used Components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 2.2.1 Grounded-sidewall-shielded microstrip line . . . . . . . . . . . . . . . . . . 12 2.2.2 Zero-impedance Transmission Line . . . . . . . . . . . . . . . . . . . . . . 15 2.2.3 Balun . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 2.2.3.1 Active Balun . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 2.2.3.2 Passive Balun . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 2.3 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 3 Low-power Low-noise Amplifiers 25 3.1 173-207 GHz Ultra-low-power Amplifier . . . . . . . . . . . . . . . . . . . . . . . 25 3.1.1 Topology Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 3.1.2 Bias Dependency of the Small-signal Performance . . . . . . . . . . . . . 27 3.1.2.1 Bias . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 3.1.2.2 Bias vs Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 3.1.2.3 Bias vs Noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 3.1.2.4 Bias vs Stability . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 3.1.3 Bias selection and Device sizing . . . . . . . . . . . . . . . . . . . . . . . . 36 3.1.3.1 Bias Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 3.1.3.2 Device Sizing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37 3.1.4 Performance Enhancement Technologies . . . . . . . . . . . . . . . . . . . 41 3.1.4.1 Gm-boosting Inductors . . . . . . . . . . . . . . . . . . . . . . . 41 3.1.4.2 Stability Enhancement . . . . . . . . . . . . . . . . . . . . . . . 43 3.1.4.3 Noise Improvement . . . . . . . . . . . . . . . . . . . . . . . . . 45 3.1.5 Circuit Realization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 3.1.5.1 Layout Scheme . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 3.1.5.2 Inductors Design . . . . . . . . . . . . . . . . . . . . . . . . . . . 47 3.1.5.3 Dual-band Matching Network . . . . . . . . . . . . . . . . . . . 48 3.1.5.4 Circuit Implementation . . . . . . . . . . . . . . . . . . . . . . . 50 3.1.6 Results and Discussions . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51 3.1.6.1 Measurement Setup . . . . . . . . . . . . . . . . . . . . . . . . . 51 3.1.6.2 Measurement Results . . . . . . . . . . . . . . . . . . . . . . . . 51 3.1.6.3 Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54 3.2 72-108 GHz Low-Power Tunable Amplifier . . . . . . . . . . . . . . . . . . . . . . 55 3.2.1 Configuration, Sizing, and Bias Tuning Range . . . . . . . . . . . . . . . . 55 3.2.2 Regional Matching Network . . . . . . . . . . . . . . . . . . . . . . . . . . 57 3.2.2.1 Impedance Variation . . . . . . . . . . . . . . . . . . . . . . . . . 57 3.2.2.2 Regional Matching Network Design . . . . . . . . . . . . . . . . 60 3.2.3 Circuit Implementation . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65 3.2.4 Results and Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65 3.2.4.1 Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65 3.2.4.2 Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68 3.3 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71 4 Low-power Down-conversion Mixers 73 4.1 97 GHz Low-power Down-conversion Mixer . . . . . . . . . . . . . . . . . . . . . 74 4.1.1 Mixer Design and Implementation . . . . . . . . . . . . . . . . . . . . . . 74 4.1.1.1 Mixer Topology . . . . . . . . . . . . . . . . . . . . . . . . . . . 74 4.1.1.2 Bias Selection and Device Sizing . . . . . . . . . . . . . . . . . . 77 4.1.1.3 Mixer Implementation . . . . . . . . . . . . . . . . . . . . . . . . 79 4.1.2 Results and Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80 4.1.2.1 Measurement Results . . . . . . . . . . . . . . . . . . . . . . . . 80 4.1.2.2 Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82 4.2 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83 5 Low-power Multipliers 87 5.1 General Design Flow . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88 5.2 56-66 GHz Low-power Frequency Quadrupler . . . . . . . . . . . . . . . . . . . . 89 5.3 172-201 GHz Low-power Frequency Tripler . . . . . . . . . . . . . . . . . . . . . 93 5.4 176-193 GHz Low-power ×12 Frequency Multiplier . . . . . . . . . . . . . . . . . 96 5.5 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100 6 Low-power Receivers 101 6.1 Receiver Performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101 6.2 LO Chain (×12) Integrated 190 GHz Low-Power Receiver . . . . . . . . . . . . . 104 6.2.1 Receiver Architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 105 6.2.2 Low-power Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . 107 6.2.3 Building Blocks . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 108 6.2.3.1 LNA and LO DA . . . . . . . . . . . . . . . . . . . . . . . . . . 108 6.2.3.2 Tunable Mixer and IF BA . . . . . . . . . . . . . . . . . . . . . 111 6.2.3.3 65 GHz (V-band) Quadrupler . . . . . . . . . . . . . . . . . . . 116 6.2.3.4 G-band Tripler . . . . . . . . . . . . . . . . . . . . . . . . . . . . 120 6.2.4 Receiver Results and Discussion . . . . . . . . . . . . . . . . . . . . . . . 123 6.2.5 Measurement Setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124 6.2.6 Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124 6.3 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 131 7 Conclusions 133 7.1 Summaries . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 133 7.2 Outlook . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 134 Bibliography 135 List of Figures 149 List of Tables 157 A Derivation of the Gm 159 A.1 Gm of standard cascode stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . 159 A.2 Gm of cascode stage with Lcas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 160 A.3 Gm of cascode stage with Lb . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 161 B Derivation of Yin in the stability analysis 163 C Derivation of Zin and Zout 165 C.1 Zin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 165 C.2 Zout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 167 D Derivation of the cascaded oP1dB 169 E Table of element values for the designed circuits 171

Nanophotonische Systeme sind eine leistungsfähige Plattform für die Untersuchung von Licht-Materie-Wechselwirkungen. In solchen Systemen bricht die übliche Beschreibung einer elektromagnetischen Welle als eine Welle, die in Bezug auf ihre Ausbreitungsrichtung transversal polarisiert ist, zusammen. Dies ist auf die Einengung der geführten Lichtfelder zurückzuführen, welche zu einer longitudinalen Komponente der elektromagnetischen Felder führt. In dieser Arbeit nutzen wir dies in Verbindung mit unterschiedlichen Kopplungsstärken von Cäsiumatomen an \sigma^- und \sigma^+ polarisiertes Licht, um das Prinzip neuartiger nicht-reziproker optischer Bauelemente zu demonstrieren. Im ersten Teil dieser Arbeit demonstrieren wir die nicht-reziproke Verstärkung von fasergeführtem Licht mit Hilfe von Raman-Verstärkung durch spinpolarisierte Cäsiumatome, die an die Nanofasertaille eines verjüngten Faserabschnitts gekoppelt sind. Wir zeigen, dass unser neuartiger Mechanismus kein externes Magnetfeld benötigt und dass wir die Richtung der Verstärkung vollständig über den atomaren Spinzustand kontrollieren können. Darüber hinaus nutzen wir die chirale Licht-Materie-Wechselwirkung in unserem System, um einen nicht-reziproken antisymmetrischen optischen Phasenschieber zu realisieren. Diese Ergebnisse tragen zur Etablierung einer neuen Klasse von spin-gesteuerten, nicht-reziproken integrierten optischen Bauelementen bei und können den Aufbau komplexer optischer Netzwerke vereinfachen. In einem weiteren Forschungsprojekt tragen wir zum grundlegenden Verständnis von Atomen bei, indem wir die Lebensdauer eines angeregten Cäsiumzustands präzise messen. Wir messen die Lebensdauer des Cäsium 5D_{5/2} Zustands im freien Raum. Wir finden eine Lebensdauer von 1353(5) ns, die mit einer aktuellen theoretischen Vorhersage übereinstimmt. Unsere Messung trägt dazu bei, eine seit langem bestehende Unstimmigkeit zwischen verschiedenen experimentellen und theoretischen Ergebnissen zu beseitigen. / Nanophotonic systems are a powerful platform for the study of light-matter interactions. In such systems, the common description of an electromagnetic wave as a wave that is transversely polarized with respect to its propagation direction breaks down. This is due to the tight confinement of the guided light fields, which leads to a longitudinal component of the electromagnetic fields. In this thesis, we use this in conjunction with different coupling strengths of cesium atoms to \sigma^- and \sigma^+ polarized light to provide proof-of-principle demonstrations of novel non-reciprocal optical devices. In the first part of this thesis, we demonstrate non-reciprocal amplification of fiber-guided light using Raman gain provided by spin-polarized cesium atoms that are coupled to the nanofiber waist of a tapered fiber section. We show that our novel mechanism does not require an external magnetic field and that it allows us to fully control the direction of amplification via the atomic spin state. Moreover, we use the chiral light-matter interaction in our system to implement a non-reciprocal antisymmetric optical phase shifter. These results contribute to establishing a new class of spin-controlled, non-reciprocal integrated optical devices and may simplify the construction of complex optical networks. In an additional research project, we also contribute to the fundamental understanding of atoms by precisely measuring the lifetime of an excited cesium state. We measure the lifetime of the cesium 5D_{5/2} state in free space. We find a lifetime of 1353(5) ns, in agreement with a recent theoretical prediction. Our measurement contributes to resolving a long-standing disagreement between several experimental and theoretical results.

This work explores the application of various power amplifier design techniques for mobile communications. Several circuit configurations including class A amplifier, Doherty amplifier and power combining amplifier have been developed, which are to improve the performance of power amplifiers in terms of power added efficiency transmission power and bandwidth. In chapter 2, the cascode PA adopting tuning capacitor structure is proposed and implemented to enhance the efficiency. In chapter 3, a novel Doherty amplifier configuration using a 3-stage polyphase filter as power splitter is introduced. Moreover, the second harmonic cancellation function of balun combining PA is analysed and verified with experimental results in chapter 4. The fully integrated cascode class A amplifier adopts RC negative feedback, which is to enhance bandwidth and input/output matching. The integrated choke inductor compensating the parasitic capacitor of transistors has very low quality factor, which decreases the efficiency of the power amplifier. To reduce the inductance value of the choke inductor, a tuning capacitor is connected in parallel with the choke inductor. As a result, the inductor resistance is reduced as well, which diminishes the power consumption induced by the resistance of the choke inductor. This proposed PA configuration is validated by simulation results with the PAE improved by 3 % at the 1 dB compression point compared to the topology without tuning capacitor. The experimental results demonstrate a PA which delivers an output power of 21.3 dBm with PAE of 21 % at the 1 dB compression point. The Doherty amplifier with 2-way Wilkinson power splitter is integrated in a 0.9 mm×1.8 mm chip. The main and peak amplifier adopt cascode configuration to improve the stability of the Doherty amplifier. To minimize the chip size, the quarter wave transmission line in the topology is replaced by π-type lumped element equivalent network. To increase the operating bandwidth, the Doherty amplifier configuration using a 3-stage polyphase filter as power splitter is proposed. The topology consists of 3-stage RC polyphase filter, drive amplifiers, main amplifier, peak amplifier, and impedance inverter. By employing the polyphase filter, the quarter-wave transmission line at the input of the peak amplifier for compensating the phase shift of the impedance inverter is eliminated. According to the analysis of the polyphase filter prototype, the 3-stage polyphase filter is selected, and the component parameters are determined. The main amplifier and peak amplifier are using differential cascode configuration. The drive amplifier is to increase the power gain and provide proper impedance matching for the Doherty amplifier. The results demonstrate an outstanding broadband Doherty amplifier with a bandwidth of 1.8 GHz. The chip temperature rises dramatically due to the high power consumption of power amplifier. Consequently, the collector currents of the SiGe transistors are varying with the changing temperature, which deteriorates the PA performance. In the improved 3-stage PPF Doherty design, the bias voltages of the transistors in the first version 3-stage PPF Doherty amplifier are replaced by reference currents feeding through bias circuits. With current sources providing bias current to the transistors, the performance of the improved Doherty amplifier is enhanced. The power combining PAs are constructed on FR-4 PCB boards using discrete components. The single ended power amplifier in the power combining PA is built with high linearity HEMT transistor. The balun combining PA has an advantage of second harmonic cancellation, which is validated by both analysis and measurements. Moreover, power combining PAs with 2-way transmission line and lumped element Wilkinson power divider are designed. The transmission lines in these designs are analyzed using EM simulation tool and verified with testing structures on PCB boards.

Die Modulation von Distorsionsprodukt-Otoemissionen (DPOAE) durch Töne tiefer Frequenz ist ein Methode, mit der die Auswirkung von Verlagerungen der kochleären Trennwand auf die Funktion des kochleären Verstärkers untersucht werden kann. Damit bietet sie einen neuen objektiven Ansatz zur Diagnostik unterschiedlicher kochleärer Hörstörungen sowie zur Untersuchung physiologischer Mechanismen der Kochlea. Durch einen tieffrequenten Ton (f = 32,8 Hz) werden die DPOAE in Abhängigkeit von dessen Phase periodisch verändert. Die Ausprägung der Modulation hängt von den Parametern des Tieftons und der Primärtöne (mit den Frequenzen f1 und f2) ab. Bei zwölf normalhörenden Probanden wird der Einfluß des Tieftonpegels, der Primärtonpegel und der Primärtonfrequenzen auf die Modulation der DPOAE mit der Frequenz 2f1-f2 untersucht. Die Ergebnisse dieser Methode werden mit denen der subjektiven Phasenaudiometrie bei diesen Probanden verglichen. Mit den Primärtonfrequenzen f1 = 2,5 und f2 = 3 kHz steigt die mittlere Modulationstiefe der DPOAE mit zunehmendem Tieftonpegel sowie mit abnehmenden Primärtonpegeln nichtlinear. Mit hohem Tieftonpegel (L = 115 dB SPL) und geringen Primärtonpegeln (bis zu L1 = 50 und L2 = 30 dB HL) sind DPOAE-Pegelverläufe mit zwei Minima und zwei Maxima pro Tieftonperiode zu beobachten. Die Pegelminima liegen kurz nach der maximalen Druck- bzw. Sogphase des Tieftons vor dem Trommelfell, entsprechend der maximalen Auslenkung der kochleären Trennwand in Richtung Scala vestibuli bzw. Scala tympani. Sie zeigen eine mittlere Latenz von 4 ms gegenüber den Verdeckungsmaxima der subjektiven Mithörschwelle im Phasenaudiogramm, die wahrscheinlich durch die Summe der Antwortzeit der aktiven kochleären Prozesse und der Laufzeit der DPOAE-Signale retrograd aus der Kochlea zur Meßsonde im äußeren Gehörgang zustande kommt. Mit geringeren Tieftonpegeln (L = 110 dB SPL) bzw. höheren Primärtonpegeln (ab L1 = 55 und L2 = 40 dB HL) sowie höheren Primärtonfrequenzen (f1 = 4, f2 = 4,8 kHz) sind nur noch jeweils ein DPOAE-Pegelminimum und -maximum pro Tieftonperiode zu beobachten. Mit den Primärtonfrequenzen f1 = 5 und f2 = 6 kHz zeigt sich keine Modulation der DPOAE mehr. Die Ergebnisse werden unter Verwendung einer Boltzmannfunktion zweiter Ordnung als Annäherung an die mechano-elektrische Transferfunktion äußerer Haarzellen simuliert. Bei einigen Probanden werden außerdem die Modulation der DPOAE mit der Frequenz 3f1-2f2 durch den tieffrequenten Ton sowie der Einfluß spontaner otoakustischer Emissionen (SOAE) auf die Messung tieftonmodulierter DPOAE untersucht. / Low-frequency modulation of distortion product otoacoustic emissions (DPOAE) is a method which allows to investigate the effect of the displacement of the cochlear partition on the function of the active cochlear process. It offers a new objective approach to diagnose different sensory hearing disorders as well as to investigate physiological cochlear mechanisms. The DPOAE are modulated by a low-frequency tone (with the frequency f = 32,8 Hz), depending on its phase. The extent of this modulation depends on the acoustic parameters of the suppressing low-frequency tone and the stimulating primary tones (f1 and f2). In twelve normal hearing subjects the influence of the low-frequency tone level, the levels and the frequencies of the primary tones on the modulation of the DPOAE with the frequency 2f1-f2 are investigated. In these subjects, the phase-dependent masked subjective threshold is also registered. The results of both methods are compared. With the primary tone frequencies f1 = 2,5 and f2 = 3 kHz the mean value of the DPOAE modulation depth presents a nonlinear growth with increasing low-frequency tone level and decreasing primary tone levels, respectively. With high low-frequency tone level (L = 115 dB SPL) and low primary tone levels (up to L1 = 55 and L2 = 40 dB HL), the time course of the DPOAE level shows two minima and two maxima within one period of the low-frequency tone. The minimal DPOAE levels are registered shortly after the phases of maximal condensation and rarefaction of the low-frequency tone in front of the eardrum, respectively, corresponding to the largest displacement of the cochlear partition towards the scala tympani and the scala vestibuli. The time course of the DPOAE level shows a mean latency of 4 ms with regard to the masking patterns of the phase-dependent masked threshold, due to the response time of the active cochlear process and the retrograde travelling time of the DPOAE. With lower low-frequency tone levels (L1 = 110 dB SPL), higher primary tone levels (from L1 = 55, L2 = 40 dB HL), and higher primary tone frequencies (f1 = 4, f2 = 4,8 kHz), respectively, the DPOAE level presents only one maximum and one minimum per period of the low-frequency tone. With the primary frequencies f1 = 5 and f2 = 6 kHz no modulation of the DPOAE is registered. The results are simulated using a second-order Boltzmann function as an approximation of the mechano-electric transfer function of the outer hair cells. Additionally, in some subjects the low-frequency modulation of the DPOAE with the frequency 3f1-2f2 and the influence of spontaneous otoacoustic emissions (SOAE) on the registration of low-frequency modulated DPOAE are investigated.

This paper presents an overview of the research work currently being performed within the frame of project DAAB and its successor DAAB-TX towards the integration of ultra-wideband transceivers operating at mm-wave frequencies and capable of data rates up to 100 Gbits–¹. Two basic systemarchitectures are being considered: integrating a broadband antenna with a distributed amplifier and integrate antennas centered at adjacent frequencies with broadband active combiners or dividers. The paper discusses in detail the design of such systems and their components, fromthe distributed amplifiers and combiners, to the broadband silicon antennas and their single-chip integration. All components are designed for fabrication in a commercially available SiGe:C BiCMOS technology. The presented results represent the state of the art in their respective areas: 170 GHz is the highest reported bandwidth for distributed amplifiers integrated in Silicon; 89 GHz is the widest reported bandwidth for integrated-system antennas; the simulated performance of the two antenna integrated receiver spans 105 GHz centered at 148GHz, which would improve the state of the art by a factor in excess of 4 even against III-V implementations, if confirmed by measurements.

Die vorliegende Arbeit behandelt integrierte Hochvolt-Schaltungen für die Ansteuerung elektrostatisch arbeitender Mikroaktoren und Mikroaktorarrays. Im Besonderen wird auf die Gesichtspunkte der Treiberschaltungen von Torsionsspiegelarrays eingegangen. Es werden verschiedene Verstärkerbetriebsarten und Schaltungsvarianten hinsichtlich der Ansteuerung kleiner kapazitiver Lasten beurteilt. Für die hocheffiziente Signalübertragung zwischen Low-Side und High-Side in geschalteten Hochvolt-Verstärkern wird ein neuer dynamischer Level-Shifter vorgestellt. Anhand eines gebondeten Mikroelektronik-Mikromechanik-Aufbaus für ein Hadamard-Transformations-Spektrometer werden die speziellen Aspekte des Elektronikentwurfs für ein System-in-Package aufgezeigt. Als Entwurfsgrundlage wird ein Überblick über die wesentlichen Isolationstechnologien für integrierte Hochvolt-Schaltungen und über die Bauelementemodellierung in einer SOI-Technologie ausgearbeitet. Außerdem werden die Vor- und Nachteile der wichtigsten Antriebsprinzipien von Mikroaktoren zusammengefasst.

This work was performed in the framework of two application-oriented research projects that focus on the generation and evaluation of fluorescent Saccharomyces (S.) cerevisiae-based sensor and reporter cells for white biotechnology as well as the extension of the conventional single-cell/single-construct principle of ordinary yeast biosensor approaches. Numerous products are currently generated by biotechnological processes which require continuous and precise process control and monitoring. These demands are only partially met by physical or physiochemical sensors since they measure parameters off-line or use surrogate parameters that consequently provide only indirect information about the actual process performance. Biosensors, in particular whole cell-based biosensors, have the unique potential to near-line and long-term monitor parameters such as nutrient availability during fermentation processes. Moreover, they allow for the assessment of an analyte’s biological relevance. Prototype yeast sensor and reporter strains derived from common laboratory strains were transformed with multicopy expression plasmids that mediate constitutive or inducible expression of a fluorescence reporter gene. Performance of these cells was examined by various qualitative and quantitative detection methods – representative of putative transducer technologies. Analyses were performed on the population level by microplate reader-based fluorometry and Western blot as well as on the single-cell level by fluorescence microscopy and flow cytometry. ‘Signature’ promoters that are activated or repressed during particular nutrient-limited growth conditions were selected in order to generate yeast nutrient sensor strains for monitoring the biological availability of nitrogen, phosphorus or sulphur. For each category, at least one promoter mediating at least threefold changed green fluorescence levels between sensor cells in non-limited and nutrient-limited conditions was identified. Sensor strains were evaluated in detail regarding sensitivity, analyte selectivity and the ability to restore basic fluorescence after shift from nutrient-limited to non-limited conditions (regeneration). The applicability for bioprocess monitoring purposes was tested by growth of yeast nutrient sensor cells in microalgae media and supernatants. Despite successful proof of principle, numerous challenges still need to be solved to realise prospective implementation in this field of white biotechnology. The major drawback of plasmid-borne detection constructs is a high fluorescence variance between individual cells. By generation of a nitrogen sensor strain with a genome-integrated detection construct, uniform expression on the single-cell level and simultaneous maintenance of basic properties (ability of fluorescence induction/regeneration and lack of cross-reactivity) was achieved. However, due to the singular detection construct per cell, significantly weaker overall fluorescence was observed. The traditional single-cell/single-construct approach was expanded upon in two ways. Firstly, a practical dual-colour sensor strain was created by simultaneous, constitutive expression of a red fluorescence reporter gene in green fluorescent nitrogen sensor cells. Secondly, an innovative cellular communication and signal amplification system inspired by the natural S. cerevisiae pheromone system and mating response was established successfully. It features the yeast pheromone alpha-factor as a trigger and alpha-factor-responsive reporter cells which express a fluorescence reporter gene from the pheromone-inducible FIG1 promoter as an output signal. The system was functional both with synthetic and cell-secreted alpha-factor, provided that recombinant cells were deleted for the alpha-factor protease Bar1p. Integration of amplifier cells which secrete alpha-factor in response to stimulation with the pheromone itself could increase the system\'s sensitivity further. Signal amplification was demonstrated for phosphorus sensor cells as a proof of concept. Therefore, the alpha-factor-based cellular communication and signal amplification system might be useful in applications that suffer from poor signal yield. Due to its modular design, the system could be applied in basically any cell-based biosensor or sensor-actor system. Immobilisation of the generated sensor and reporter cells in transparent natural polymers can be beneficial considering biosensor fabrication. Functionality of sensor and reporter cells in calcium-alginate beads or nano-printed arrays was successfully demonstrated. For the latter setup, fluorescence scanning and software-assisted fluorescence quantification was applied as a new detection method. In an experiment using an agarose-based two-compartment setup proposed by Jahn, 2011, properties of the alpha-factor-based cellular communication and signal amplification system after immobilisation were tested. These studies provide an initial experimental basis for an appropriate geometry of miniaturised immobilisation matrices with fluorescent yeast sensor and reporter cells in prospective biosensor designs.

A K-band integrated super-regenerative amplifier (SRA) in a 130nm SiGe BiCMOS technology is designed and characterized. The circuit is based on a novel stacked transistor differential cross-coupled oscillator topology, with a controllable tail current for quenching the oscillations. The fabricated integrated circuit (IC) occupies an area of 0.63mm2, and operates at the free-running center frequency of 25.3 GHz. Characterization results show circuit operation from a minimum input power level required for a phase coherent output as −110 dBm, and the input power level corresponding to the linear to logarithmic mode transition of −85 dBm, the lowest reported for K-band integrated logarithmic mode SRAs to date to the knowledge of the authors. The measured output power is 7.8dBm into a 100 differential load. The power consumption of the circuit is 110mW with no quench signal applied, and 38mW with 30 % duty cycle quenching. The quench waveform designed for the reported measurement result is also discussed.

This letter presents a low-noise amplifier with a 3 dB-bandwidth, from 55 to 74 GHz, excellent noise performance and low power consumption based on a three-stage common-source topology. For the first time to the authors’ best knowledge, an analytical equation that also considers the gate–drain capacitance is derived for the employed shunt–series transformer feedback input matching network. To enable shunt–series transformer feedback matching without significant gain reduction a highly asymmetric transformer is designed. Furthermore, a compact transformer-implemented T-shaped output matching network is investigated to minimize the required area. To prove these concepts, the circuit has been fabricated in a 22 nm fully depleted silicon-on-insulator technology. Thanks to the transformer-based matching, an ultra-compact active footprint of 0.039 mm² is achieved. At a power consumption of 8.4 mW from a 0.41 V supply an average noise figure of 4.8 dB and a peak gain of 14.2 dB has been measured. In- and output matching better than −10 dB over the 19 GHz wide 3 dB-bandwidth are demonstrated.