For III-nitride wide-bandgap materials, the lack of efficient p-type wide bandgap semiconductors limits the full potential of group-III nitride-based optoelectronic devices. Conventional wide bandgap p-type materials consisting of magnesium-doped gallium nitride (GaN:Mg) and magnesium-doped aluminum gallium nitride (AlGaN:Mg) typically exhibit low hole carrier concentrations of <1018 cm-3 . Hence, I used different wide bandgap inorganic p-type materials as a promising solution, e.g., copper thiocyanate (CuSCN). CuSCN has multiple attractive properties that hold potential for applications in III-nitride materials. For example, its energy band gap is up to 3.9 e.V and its electron effective mass is higher than its hole effective mass. These two key features make CuSCN a potential wide bandgap p-type material for III-nitride systems. By exposing CuSCN to chlorine, Cl2-infused CuSCN thin film achieves a hole concentration up to 3 × 1018 cm-3 and maintains its visible-light-blind optical properties. Based on these desirable features, p CuSCN/n-GaN heterojunction ultraviolet photodetectors, as well as the p-CuSCN and n GaN interface, were fabricated to investigate the potential applications of p-CuSCN in III nitride devices. Moreover, p-CuSCN also benefits the corresponding organic solar cells; p CuSCN-based organic solar cells perform better in power conversion efficiency and stability tests under various conditions than intrinsic CuSCN-based organic solar cells. This work on p-CuSCN not only paves the way for new III-nitride semiconductor devices, but may also potentially enable the development of organic devices with better performance and longer lifetime. To explore the potential of transition metal oxides in UV photodetectors, NiO was selected to proceed with device fabrication because of its wider energy bandgap and lower hole effective mass than other transition metal oxides. Since single crystal quality is required to maintain its visible-light-blind optical property, brand-new templates were invented to grow single-crystal NiO thin films, TiN/MgO, and TiN/Si. Use of TiN thin film between NiO and the substrates provides a good back-side metal contact for NiO-based semiconductor devices. Several tools were employed to ascertain the single-crystal quality of as-grown NiO thin films on TiN/MgO and TiN/Si. I demonstrate NiO/TiN/MgO and NiO/TiN/Si bilayer structures may pave the way towards better NiO-based ultraviolet optoelectronic devices.

In the application of power inverters, power density has become a key design specification where it has stringent requirements on system size and weight. Achieving high power density need to combine lasted wide bandgap (WBG) device technology and high switching frequency to reduce passive filter size thus further shrink overall space. While still maintaining decent power conversion efficiency and low electromagnetic interference (EMI) with higher switching frequency, soft-switching needs to be implemented. A novel auxiliary resonant snubber is introduced. The design and operation are carried out, in which this snubber circuitry enables main Gallium Nitride (GaN) switches operating under zero voltage switching (ZVS) condition, and auxiliary Silicon Carbide (SiC) diodes switching under zero current switching (ZCS) condition. Besides, the auxiliary snubber circuitry gating algorithm is also optimized which allows reduction of the switching and conduction loss in auxiliary GaN switches to obtain higher system efficiency and better thermal performance. Here, this novel auxiliary resonant snubber circuitry is applied to a traditional full bridge inverter with flexible modulation suitability. This proposed inverter can be applied to a wide range of potential applications, such as string solar inverter, renewable energy combined distributed generation, dc-ac part of bi-directional electrical vehicle (EV) on-board charger, and uninterruptible power supply (UPS), etc. / Master of Science

Master of Science / Department of Electrical and Computer Engineering / Behrooz Mirafzal / Wide bandgap devices have gained increasing attention in the market of power electronics for their ability to perform even in harsh environments. The high voltage blocking and high temperature withstanding capabilities make them outperform existing Silicon devices. They are expected to find places in future traction systems, electric vehicles, LED lightning and renewable energy engineering systems. In spite of several other advantages later mentioned in this paper, WBG devices also face a few challenges which need to be addressed before they can be applied in large scale in industries. Electromagnetic interference and new requirements in packaging methods are some of the challenges being faced by WBG devices. After the commercialization of these devices, many experiments are being carried out to understand and validate their abilities and drawbacks. This paper summarizes the experimental results of various applications of mainly Silicon Carbide (SiC) and Gallium Nitride (GaN) power devices and also includes a section explaining the current challenges for their employment and improvements being made to overcome them.

GaN RF switches are widely used in today’s communication systems. With digital communications getting more and more popular nowadays, the need for improving the performance of involved RF switches is inevitable. Designing low ON-state resistance GaN switches are exceedingly important to improve the switch insertion loss, isolation and power loss. Moreover, considerations need to be taken into account to improve the switching speed of the involved GaN HEMTs. In this dissertation, a new GaN HEMT structure called “Tunnel MOS Heterostructure FET (TMOSHFET)” is introduced which has lower ON-state resistance and faster switching speed compared to conventional AlGaN/GaN HEMTs. In the switch ON process, the channel of this device is charged up by electron tunneling from a layer underneath the channel as opposed to typical AlGaN/GaN HEMTs in which electron injection from the source is charging up the channel. The tunneling nature of this process together with the shorter travel distance of electrons in TMOSHFET provide for a faster switching speed. In order to understand the tunneling mechanisms in TMOSHFET, the fabrication of AlGaN/GaN Schottky Barrier Diodes (SBDs) with various AlGaN thicknesses is demonstrated on Si (111) substrate. The impacts of SF6 dry etching on the trap density and trap state energy of AlGaN surface are investigated using the GP/w- w method. Various tunneling mechanisms at different biases are then characterized in samples and compared with each other. To improve the source and drain resistances in TMOSHFET, a model is generated to optimize the 2DEG density and electric field in AlGaN/GaN heterostructure based on Al mole fraction, AlGaN thickness and the thickness of SiN passivation layer and it is experimentally verified by non-contact Hall 2DEG density measurements. The spontaneous and piezoelectric polarizations together with strain relaxation have been implemented into the model, taking into account the annealing effects. From the experimental data on obtained parameters, the operation and device parameterization of the TMOSHFET is outlined and design considerations to improve the device R_(ON)-V_(BR) figure of merit are discussed.

No description available.

No description available.

Because of the increasing emphasis on environmental concerns, there has been a growing demand for lower fuel consumption in modern transportation applications. To reduce fuel comsumption, higher efficiency, higher power density power converters are desired. The new generation of wide bandgap (WBG) power semiconductor devices pushs the switching frequency and output power of the electric system in transportation to a higher level thanks to their higher blocking voltage, higher operating frequency, and smaller parasitic elements. With benefits such as size reudcetion, costs saving, and reliability improvement, integration technologies have been widely adopted in power electronic systems, especially with the emergence of WBG semiconductor devices. These improvements will futher translate into reduced fuel consumption, extended operating range, and increased passenger compartment. Transportation applications pose a challenging environment for converter integration. The fast switching speed and the high blocking voltage of WBG semiconductor devices also put forward higher requirements for converter integration. First, the power converters used in transportation applications are often powered from the batteries that support multiple loads. During load changes, crank, or jump-start, undesired transients exist, which requires the power converters to be capable of operating under a wide-input-voltage range. This requirement results in a very limited design region of acceptance, making the converter hard to handle uncertainties. However, the integration process might bring large uncertainties, such as material property changes. This phenomenon can degrade converter performance or even cause design failures. Besides, the power converters for transporation applications often work in harsh environment, such as high ambient temperature or low air density. The former can lead to overheated and the latter degrades insulation strength, both of which hinder high power density design. Moreover, with the advent of all kinds of portable devices, converters are required to deliver more power. The introduction of universal serial bus (USB) power delivery (PD) extends the delivered power. To meet the specification, the power converters should provide a wide-output-voltage range, which brings challenges to the converter design. Furthermore, the charger is usually fed by an ac voltage of more than 100 V, which is then stepped down to 5 V – 20 V. The high step-down ratio increases the converter loss. To address the wide-input-voltage and high-temperature challenges, a dual-output, PCB-embedded transformer based active-clamp Flyback (ACF) gate-drive power supply (GDPS) for automotive applications is proposed. It has been demonstrated that the PCB-embedding technique effectively improves converter power density. The final prototype achieves a power density of 53.2 W/in3, a peak efficiency of 89.7 %, a transformer input-output capacitance of 9.7 pF, an input-voltage range of 9.9 V – 28 V, and a maximum operating temperature at low-line (LL) voltage of 105 °C and 115 °C at high-line (HL) voltage. Yet the above unit failed to meet all of the design targets due to the material property degradation in transformer. This degradation is caused by the mechanical stress induced in the integration process. To investigate its impact on wide-input-voltage converter design, several PCB-embedded magnetic boards are fabricated with different core materials and stress levels. Based on the analysis, experimentally derived correction factors are proposed and applied to the models used in the multi-objective optimization (MDO) process. The improved design successfully achieves the targeted wide-input-votlage range. When aircrafts climb during flight, air density reduces and the breakdown voltage decreases correspondingly. The insulation design becomes a challenge for the gate driver for SiC-based airborne applications. To provide sufficient insulation strength and achieve high power density simultaneously, a Paschen curve based insulation co-ordination is proposed. Electric-field control methodology is applied to the layout design. By properly designing the field control plates, the peak electric field has been shifted from the air to fr4 material that features much higher dielectric strength. The proposed gate driver attains a small size of 128.7 mm × 61.2 mm × 23.8 mm. Partial discharging tests are conducted in an altitude chamber. The experimental result shows that the proposed gate driver provides sufficient insulation strength at 50, 000 ft. To tackle the wide-output-voltage range and high-step-down ratio challenges in the USB-C PD charger in airborne applications, a LLC converter with PCB-winding based transformer with built-in leakage inductance is presented. A flying-capacitor based voltage divider (FCVD) switching bridge is proposed to replace the conventional half-bridge or full-bridge switching bridge. The propsed FCVD shows a current reduction of over 50 % than the conventional half-bridge with the same circuit elements. The prototype achieves a high efficiency of 90.3 % to 93.2 % over 5 V to 20 V outputs, and a high power density of 73.2 W/ in³, which is almost two time larger than the state-of-the-art power density. Partial discharging tests are also conducted in an altitude chamber. A partial discharing inspection voltage of 800 V is found at 10, 000 ft, which is much higher than the requirement. / Doctor of Philosophy / Because of the increasing emphasis on environmental concerns, there has been a growing demand for lower fuel consumption in modern transportation applications. The new generation of wide bandgap (WBG) power semiconductor devices and various integration technologies enable electronic systems in transportation to achieve higher efficiency and higher power density. These improvement will futher translate into reduced fuel consumption, extended operating range, and increased passenger compartment. However, transportation applications put more requirements on power converter designs. This dissertation, therefore, focusing on addressing the integration challenges in high power density WBG-based circuits for transportation applications from the aspects of wide-input-voltage range, material properties degradation, harsh environment, and wide-output-voltage range together with high step-down ratio. To meet the wide-input-voltage and high temperature requirements in automotive applications, a dual-output, PCB-embedded transformer based active-clamp Flyback (ACF) dc-dc converter is proposed. The final prototype achieves a power density of 53.2 W/in3, a peak efficiency of 89.7 %, a transformer input-output capacitance of 9.7 pF, an input-voltage range of 9.7 V â€" 28 V, and a maximum operating temperature at low-line (LL) voltage of 105 °C and 115 °C at high-line (HL) voltage. Yet the above unit failed to meet all of the design targets due to the material property degration in PCB-embedded transformer. This degradation is caused by the mechanical stress during integration process. To investigate its impact on automotive converter, several PCB-embedded magnetic boards are fabricated with different core materials and stress levels. Based on the analysis, experimentally derived correction factors are proposed and applied to the models used in the multiobjective optimization process. The improved design successfully achieves the targeted wide-input-votlage range. When aircrafts climb during flight, air density reduces and thus insulation strength decreases correspondingly. Instead of using oversized altitude correction factors provided by IEC standards, a Paschen curve based insulation co-ordination is proposed. Electric-field control methodology is applied to the gate driver layout. The proposed gate driver attains a small size of 128.7 mm × 61.2 mm × 23.8 mm. Partial discharging test is conducted in an altitude chamber. The experimental result shows that the proposed gate driver provide sufficient insulation strength at 50, 000 ft. To tackle the wide-output-voltage range and high-step-down ratio challenges in the USB-C PD charger in airborne applications, a LLC converter with PCB-winding based transformer with built-in leakage inductance is presented. A flying-capacitor-based voltage divider (FCVD) switching bridge is proposed to replace the conventional half-bridge or full-bridge switching bridge. The propsed FCVD shows a current reduction of over 50 % than the conventional half-bridge with the same circuit design. The prototype achieves a high efficiency of 90.3 % to 93.2 % over 5 V to 20 V outputs, and a high power density of 73.2 W/ in3, which is more than two time larger than the state-of-the-art power density. Partial discharging tests are also conducted in an altitude chamber. A partial discharing inspection voltage (PDIV) of 800 V is found at 10, 000 ft, which is much higher than the requirement.

Current AlGaN/GaN Heterojunction Field Effect Transistors (HFETs) make use of multiple sources, drains, and gates in parallel to maximize transconductance and effective gain while minimizing the current density through each channel. To connect the sources to a common ground, current practice prescribes the fabrication of air bridges above the gates and drains. This practice has the advantage of a low dielectric constant and low parasitic capacitance, but it is at the expense of manufacturability and robust device operation. In the study described below, the air bridges in AlGaN/GaN HFETs were replaced by a polymer supported metallization bridge with the intention of improving ease of fabrication and reliability. The DC, high frequency, and power performance for several polymer step heights were investigated. The resultant structures were functional and robust; however, their electrical performance was degraded due to high source resistance. The cause of the high source resistance was found to be thinning of the metallization at the polymer step. The effect was more pronounced for higher step heights. / Master of Science

Wir untersuchen die Phasenbildung von Festkörperlösungen von (InxGa1-x)2O3 experimentell mittels Transmissionselektronenmikroskopie und stützen uns bei der Modellierung auf die Clusterexpansion. Epitaktische (InxGa1-x)2O3 Schichten auf kristallinen Substrate sind durch ausgeprägte Ordnung auf den Kationenuntergittern gekennzeichnet, bei welchem In und Ga sich auf Gitterplätze einbauen auf denen sie die energetisch günstigste Koordination zum Sauerstoff einnehmen. Ausgehend von diesem Befund, modifizieren wir das Modells der idealen Mischung so dass wir die Konfigurationsentropie auf den kationischen Untergittern mit spezifischer Koordinations getrennt betrachten um diese realistisch zu berechnen. Das resultierende Phasendiagramm ist durch enge thermodynamisch Stabilitätsbereiche für die jeweiligen Phasen gekennzeichnet, weil sich gleichzeitig große metastabile Zusammensetzungsbereiche ergeben bei Temperaturen die typisch für epitaktisches Wachstum sind: so ist die monokline Phase im Zusammensetzungsbereich x<0.5 metastabil, die hexagonale Phase für 0.55<x<0.7 und die kubische Bixbyit-Phase für x>0.91. Wird amorphes (InxGa1-x)2O3 kristallisiert in-situ im TEM, bildet sich im Zusammensetzungbereich bis x<0.22 die Spinellphase, die als ungeordnete Variante der monoklinen Phase beschrieben wird. Oberhalb dieser Zusammensetzung ist die kubische Phase stabil. Ursache hierfür ist der Einfluss der maximale Menge an Konfiguartionsentropie auf die Bildungsenthalpie in Strukturen mit vielfältigem Koordinationsumgebungen der Kationen. Der letzte Teil der Arbeit befasst sich mit dem Einflusses der Gitterordnung auf den Materialkontrast bei der Abbildung mittels HAADF (High Angle Annular Dark Field) STEM. Hier wird gezeigt, dass die Anregung des 2s-Bloch-Wellen-Zustands zu langperiodsichen Kontrastoszillationen führt, die die quantitaive Bestimmung der Zusammensetzung mittels Z-Kontrast erschwert es aber erlaubt den Ordnungsparameter bei bekaannter Zusammensetzung zu messen. / We investigate the phase formation in (InxGa1-x)2O3 solid solutions experimentally by means of transmission electron microscopy (TEM) and with computational support using cluster expansion. In the case of epitaxial growth on crystalline substrates, we find strong ordering on the cation sublattices of (InxGa1-x)2O3, energetically driven by the tendency of In and Ga to each assume their preferred coordination environment. Based on this experimental finding, we modify the model of the ideal mixture by considering the configurational entropy on the respective cation sublattices with different coordination separately in order to calculate it realistically. The resulting phase diagram is characterized by narrow thermodynamically stable ranges for each phase, while wide composition ranges of metastable compounds are predicted, which can be achieved at temperatures typical for epitaxy: the monoclinic phase is metastable in the composition range x<0.5, the hexagonal phase for 0.55<x<0.7, and the cubic bixbyite phase for x>0.91. If amorphous (InxGa1-x)2O3 is crystallized in-situ in the TEM, the spinel phase, which is described as a disordered variant of the monoclinic phase, is formed in the composition range up to x<0.22, while above this composition, the bixbyite phase is stable. This shift in stability is explained by the maximum amount of configurational entropy present during crystallization, which strongly influences the formation enthalpy in structures with diverse coordination environments of the cations. The last part of the work deals with the influence of the lattice order on the material contrast when imaging by HAADF (High Angle Annular Dark Field) STEM. It is shown that the excitation of the 2s-Bloch wave state leads to long-period contrast oscillations, which complicate the quantitative determination of the composition by Z-contrast but allows to quantify the order parameter for a given composition.

Stand der Technik sind IGBTs und Freilaufdioden aus Silizium (Si). Jahrzehntelange Forschung hat zu einer nahezu perfekten Technologie geführt. Jedoch werden die Fortschritte hinsichtlich der Reduzierung von Schalt- und Durchlassverlusten mit jeder neuen Generation von Si-IGBTs immer kleiner. Die anfallende Verlustleistung kann jedoch signifikant mit Leistungshalbleiter-Bauelementen aus Siliziumkarbid (SiC) und Galliumnitrid (GaN) gesenkt werden. Ziel dieser Arbeit ist es, zu untersuchen, ob und inwieweit mit diskreten SiC-Bipolartransistoren im TO-247- und SiC-Schottky-Dioden im TO-220-Gehäuse der Wirkungsgrad eines Antriebsstromrichters gesteigert werden kann. Ein Exkurs in die Siliziumkarbid-Halbleitertechnologie am Anfang soll deren Vorteile in Hinblick auf verlustärmere Leistungselektronik aufzeigen. Die Vorteile des Halbleitermaterials Siliziumkarbid werden anhand des SiC-Bipolartransistors im Vergleich zum ersten Leistungstransistor - dem Bipolartransistor aus Silizium - herausgearbeitet. Beim SiC-Bipolartransistor muss im laststromführenden Zustand ein Steuerstrom in die Basis eingeprägt werden. Damit erhöht sich der Treiberaufwand. Deshalb wird der erste Themenschwerpunkt auf den Treiber gelegt. In dieser Arbeit wurden ein einfacher und ein komplexer Treiber aufgebaut und evaluiert. Mit leichten Modifikationen wurden mit dem komplexeren Treiber auch IGBTs und SiC-MOSFETs für Vergleichsmessungen angesteuert. Ein neuer Ansatz zur Reduzierung der Treiberverlustleistung im Wechselrichter mit SiC-Bipolar-Transistoren wird vorgestellt. Er setzt beim Kommutierungsalgorithmus des Wechselrichters an. Ein wesentlicher Teil der Arbeit widmet sich der Charakterisierung des SiC-Bipolartransistors, insbesondere dem Schaltverhalten. Ein- und Ausschaltwärmen für verschiedene Arbeitspunkte werden ermittelt. Am Ende der Arbeit werden experimentelle Untersuchungen an einem SiC-Wechselrichter durchgeführt. Abschließend werden die Potenziale, die mit dem Einsatz von SiC-Bipolartransistoren verbunden sind, bewertet aber auch die Grenzen aufgezeigt.:1 Einleitung 1 2 Aufbau des SiC-Bipolartransistors 2.1 Siliziumkarbid (SiC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 2.1.1 Eigenschaften von monokristallinem Siliziumkarbid . . . . . . . . . . 5 2.1.2 Herstellung des SiC-Wafers . . . . . . . . . . . . . . . . . . . . . . . . 8 2.1.3 Herstellung des SiC-Bipolartransistors . . . . . . . . . . . . . . . . . . 10 2.1.4 Defekte im Siliziumkarbidkristall . . . . . . . . . . . . . . . . . . . . 11 2.2 Halbleiterphysikalische Grundlagen . . . . . . . . . . . . . . . . . . . . . . . 12 2.2.1 Gesperrter pn-Übergang . . . . . . . . . . . . . . . . . . . . . . . . . 13 2.2.2 Stromführender pn-Übergang . . . . . . . . . . . . . . . . . . . . . . . 15 2.3 Bipolartransistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 2.3.1 Aufbau und Funktionsprinzip . . . . . . . . . . . . . . . . . . . . . . . 17 2.3.2 Sperrfähigkeit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 2.3.3 Erster und zweiter Durchbruch . . . . . . . . . . . . . . . . . . . . . . 23 2.3.4 Stromverstärkung . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 2.3.5 Ladungsträgermodulation . . . . . . . . . . . . . . . . . . . . . . . . . 29 2.3.6 Eindimensionaler spezifischer Widerstand der Driftzone . . . . . . . . 30 3 Ansteuerung des SiC-Bipolartransistors 3.1 Einführung Treiber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 3.2 Herausforderungen beim Ansteuern von SiC-Bipolartransistoren . . . . . . . . 34 3.3 Treiberkonzepte für SiC-Bipolartransistoren . . . . . . . . . . . . . . . . . . . 36 3.4 Konventioneller Treiber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41 3.5 3-Level-Treiber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43 3.6 Treiber für SiC-MOSFET und IGBT . . . . . . . . . . . . . . . . . . . . . . . 45 4 Reduzierung der Treiberverluste durch Einschrittkommutierung 4.1 Einschrittkommutierung . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 4.2 Stromvorzeichenerkennung . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48 4.3 Berechnung der Verlustleistungen für den eingeschalteten Zustand des SiC- Bipolartransistors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49 4.4 Messung der Treiberverlustleistung . . . . . . . . . . . . . . . . . . . . . . . . 52 5 Charakterisierung des SiC-Bipolartransistors 5.1 Messaufbau für Untersuchung des Ein- und Ausschaltverhaltens . . . . . . . . 55 5.2 Doppelpulsverfahren . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56 5.3 Definition der Schaltzeiten und Schaltverlustleistung . . . . . . . . . . . . . . 57 5.4 Messung der Schaltwärme . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59 5.4.1 Spannungstastköpfe . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59 5.4.2 Stromsensoren . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61 5.4.3 Zeitliche Verschiebung der Messsignale . . . . . . . . . . . . . . . . . 62 5.4.4 Vergleich von konventionellem und 3-Level-Treiber . . . . . . . . . . . 65 5.4.5 Vergleich bei unterschiedlicher Treiberspannung . . . . . . . . . . . . 66 5.4.6 Vergleich bei halb und voll bestückter Halbbrücke . . . . . . . . . . . . 68 5.4.7 Vergleich von SiC-Bipolartransistor mit SiC-MOSFET und Si-IGBT . . 69 5.4.8 Reduzierung der Spannungsspitze beim Ausschalten . . . . . . . . . . 74 5.5 Simulation des Schaltverhaltens eines SiC-Bipolartransistors . . . . . . . . . . 79 5.5.1 Schaltverhalten bei Ansteuerung mit unipolarem Treiber . . . . . . . . 79 5.5.2 Simulation des Einfluss der Emitter-Induktivität auf Schaltwärme . . . 81 5.5.3 Vergleich von Simulation und Messung . . . . . . . . . . . . . . . . . 82 5.6 Durchlassverlustleistung . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85 6 Einsatz von SiC-Bipolartransistoren im Wechselrichter 6.1 Aufbau der Wechselrichter . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88 6.2 Inbetriebnahme des Wechselrichters . . . . . . . . . . . . . . . . . . . . . . . 89 6.3 Überspannungen an den Motorklemmen der 1 kW-Asynchronmaschine . . . . 91 6.4 Umbau des SiC-Wechselrichters . . . . . . . . . . . . . . . . . . . . . . . . . 93 6.5 Spannungsspitzen in der Ansteuerspannung . . . . . . . . . . . . . . . . . . . 94 6.6 Halbbrückenverluste im Leerlauf . . . . . . . . . . . . . . . . . . . . . . . . . 98 7 Zusammenfassung und Fazit 101 Literaturverzeichnis 104 A Anhang A.1 Netzliste für SiC-Bipolartransistor FSICBH057A120 . . . . . . . . . . . . . . 113 A.2 Leiterplatten für Doppelpuls-Test und SiC-Wechselrichter . . . . . . . . . . . . 114 A.3 Herleitung des Feldverlaufs in der Driftzone des gesperrten pn-Übergangs . . . 116 A.4 Herleitung des Emitterwirkungsgrads . . . . . . . . . . . . . . . . . . . . . . . 119 A.5 Herleitung des spezifischen Widerstands der Driftzone . . . . . . . . . . . . . 121 A.6 Lebenslauf von Henry Barth . . . . . . . . . . . . . . . . . . . . . . . . . . . 124 A.6.1 Persönliche Angaben . . . . . . . . . . . . . . . . . . . . . . . . . . . 124 A.6.2 Wissenschaftlicher Werdegang . . . . . . . . . . . . . . . . . . . . . . 124 / State-of-the-art are IGBTs and free-wheeling diodes made of silicon (Si). Decades of research have led to an almost perfect technology. Nevertheless, progress in terms of reduction of switching and forward conducting losses becomes smaller and smaller with each new generation of Si IGBTs. The resulting power dissipation, however, can be significantly reduced with power semiconductor devices made of silicon carbide (SiC) and gallium nitride (GaN). The objective of this work is to investigate whether and to what extent discrete SiC bipolar junction transistors (BJT) in TO-247 and SiC Schottky diodes in TO-220 packages can be used to increase the efficiency of a power drive inverter. At the beginning, a digression into silicon carbide semiconductor technology is intended to show its advantages in terms of lower-loss power electronics. The advantages of the semiconductor material silicon carbide are illustrated by the SiC bipolar junction transistor in comparison with the first power transistor - the silicon bipolar junction transistor. For the on-state of SiC bipolar junction transistors, a continuous current must be injected into the base. This increases the driving effort. Therefore, the first topic focuses on the driver. In this work, a simple and a complex driver were built and evaluated. With slight modifications, the more complex driver was also used to drive IGBTs and SiC-MOSFETs for comparative measurements. A new approach to reduce driver power dissipation in the inverter when using SiC bipolar junction transistors is presented. It focuses on the commutation algorithm of the inverter. A significant part of the work is devoted to the characterization of the SiC bipolar junction transistor, especially the switching behavior. Turn-on and turn-off switching losses for different operating points are determined. At the end of the work, experimental investigations are performed on a SiC inverter. Finally, the potentials associated with the use of SiC bipolar junction transistors are evaluated but also the limitations are shown.:1 Einleitung 1 2 Aufbau des SiC-Bipolartransistors 2.1 Siliziumkarbid (SiC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 2.1.1 Eigenschaften von monokristallinem Siliziumkarbid . . . . . . . . . . 5 2.1.2 Herstellung des SiC-Wafers . . . . . . . . . . . . . . . . . . . . . . . . 8 2.1.3 Herstellung des SiC-Bipolartransistors . . . . . . . . . . . . . . . . . . 10 2.1.4 Defekte im Siliziumkarbidkristall . . . . . . . . . . . . . . . . . . . . 11 2.2 Halbleiterphysikalische Grundlagen . . . . . . . . . . . . . . . . . . . . . . . 12 2.2.1 Gesperrter pn-Übergang . . . . . . . . . . . . . . . . . . . . . . . . . 13 2.2.2 Stromführender pn-Übergang . . . . . . . . . . . . . . . . . . . . . . . 15 2.3 Bipolartransistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 2.3.1 Aufbau und Funktionsprinzip . . . . . . . . . . . . . . . . . . . . . . . 17 2.3.2 Sperrfähigkeit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 2.3.3 Erster und zweiter Durchbruch . . . . . . . . . . . . . . . . . . . . . . 23 2.3.4 Stromverstärkung . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 2.3.5 Ladungsträgermodulation . . . . . . . . . . . . . . . . . . . . . . . . . 29 2.3.6 Eindimensionaler spezifischer Widerstand der Driftzone . . . . . . . . 30 3 Ansteuerung des SiC-Bipolartransistors 3.1 Einführung Treiber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 3.2 Herausforderungen beim Ansteuern von SiC-Bipolartransistoren . . . . . . . . 34 3.3 Treiberkonzepte für SiC-Bipolartransistoren . . . . . . . . . . . . . . . . . . . 36 3.4 Konventioneller Treiber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41 3.5 3-Level-Treiber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43 3.6 Treiber für SiC-MOSFET und IGBT . . . . . . . . . . . . . . . . . . . . . . . 45 4 Reduzierung der Treiberverluste durch Einschrittkommutierung 4.1 Einschrittkommutierung . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 4.2 Stromvorzeichenerkennung . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48 4.3 Berechnung der Verlustleistungen für den eingeschalteten Zustand des SiC- Bipolartransistors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49 4.4 Messung der Treiberverlustleistung . . . . . . . . . . . . . . . . . . . . . . . . 52 5 Charakterisierung des SiC-Bipolartransistors 5.1 Messaufbau für Untersuchung des Ein- und Ausschaltverhaltens . . . . . . . . 55 5.2 Doppelpulsverfahren . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56 5.3 Definition der Schaltzeiten und Schaltverlustleistung . . . . . . . . . . . . . . 57 5.4 Messung der Schaltwärme . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59 5.4.1 Spannungstastköpfe . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59 5.4.2 Stromsensoren . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61 5.4.3 Zeitliche Verschiebung der Messsignale . . . . . . . . . . . . . . . . . 62 5.4.4 Vergleich von konventionellem und 3-Level-Treiber . . . . . . . . . . . 65 5.4.5 Vergleich bei unterschiedlicher Treiberspannung . . . . . . . . . . . . 66 5.4.6 Vergleich bei halb und voll bestückter Halbbrücke . . . . . . . . . . . . 68 5.4.7 Vergleich von SiC-Bipolartransistor mit SiC-MOSFET und Si-IGBT . . 69 5.4.8 Reduzierung der Spannungsspitze beim Ausschalten . . . . . . . . . . 74 5.5 Simulation des Schaltverhaltens eines SiC-Bipolartransistors . . . . . . . . . . 79 5.5.1 Schaltverhalten bei Ansteuerung mit unipolarem Treiber . . . . . . . . 79 5.5.2 Simulation des Einfluss der Emitter-Induktivität auf Schaltwärme . . . 81 5.5.3 Vergleich von Simulation und Messung . . . . . . . . . . . . . . . . . 82 5.6 Durchlassverlustleistung . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85 6 Einsatz von SiC-Bipolartransistoren im Wechselrichter 6.1 Aufbau der Wechselrichter . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88 6.2 Inbetriebnahme des Wechselrichters . . . . . . . . . . . . . . . . . . . . . . . 89 6.3 Überspannungen an den Motorklemmen der 1 kW-Asynchronmaschine . . . . 91 6.4 Umbau des SiC-Wechselrichters . . . . . . . . . . . . . . . . . . . . . . . . . 93 6.5 Spannungsspitzen in der Ansteuerspannung . . . . . . . . . . . . . . . . . . . 94 6.6 Halbbrückenverluste im Leerlauf . . . . . . . . . . . . . . . . . . . . . . . . . 98 7 Zusammenfassung und Fazit 101 Literaturverzeichnis 104 A Anhang A.1 Netzliste für SiC-Bipolartransistor FSICBH057A120 . . . . . . . . . . . . . . 113 A.2 Leiterplatten für Doppelpuls-Test und SiC-Wechselrichter . . . . . . . . . . . . 114 A.3 Herleitung des Feldverlaufs in der Driftzone des gesperrten pn-Übergangs . . . 116 A.4 Herleitung des Emitterwirkungsgrads . . . . . . . . . . . . . . . . . . . . . . . 119 A.5 Herleitung des spezifischen Widerstands der Driftzone . . . . . . . . . . . . . 121 A.6 Lebenslauf von Henry Barth . . . . . . . . . . . . . . . . . . . . . . . . . . . 124 A.6.1 Persönliche Angaben . . . . . . . . . . . . . . . . . . . . . . . . . . . 124 A.6.2 Wissenschaftlicher Werdegang . . . . . . . . . . . . . . . . . . . . . . 124